An integrated active Gilbert mixer typically needs a DC bias current source at the mixer output. The DC bias current source also connects to the IF amplifier input and forms a significant source of low frequency noise, degrading the signal-to-noise ratio (SNR) of the received signal.
An example of an active Gilbert mixer 102 forming part of a mixer circuit 100 is illustrated in FIG. 1. The mixing circuit 100 comprises a voltage to current converter 101 connected to a Gilbert mixer 102. The voltage to current converter 101 comprises a pair of NMOS transistors 103a, 103b having their sources connected across a resistor Rss and to a voltage supply line Vss via respective current sources 104a, 104b, each supplying a bias current Is through the transistors 103a, 103b. The gates of transistors 103a, 103b are connected to two halves of an RF differential input signal, RF+, RF−, and the drains are connected to the Gilbert mixer 102.
The Gilbert mixer 102 comprises two pairs of NMOS transistors 105a, 105b & 106a, 106b, each pair having common sources connected to a drain of a corresponding transistor 103a, 103b in the voltage to current converter 101. Gates of the pairs of transistors 105a, 105b & 106a, 106b are connected to two halves of a differential input signal LO+, LO−, with the gates connected to each half LO+, LO− of the input signal also connected together. The LO signal may be a high frequency signal that may be applied to the LO+, LO− terminals of the mixer 102 through a coupling capacitor.
Drains of the transistors 105a, 105b, 106a, 106b, are connected to provide a differential intermediate frequency output IF+, IF−.
Harmonic rejection mixers can employ a plurality of hard-switching active Gilbert mixers connected in parallel, in order to approximate a sinusoidal effective mixing waveform. Harmonic rejection mixers are advantageously used in wideband radio transceivers such as software-defined radio. The reduced harmonic response allows a saving of RF filtering and therefore better integration. A so-called mixing-DAC, comprising a programmable array of unit mixers, constitutes a flexible and programmable implementation of a harmonic rejection mixer. An example of a harmonic rejection mixer 200 is illustrated in FIG. 2.
The mixer in FIG. 2 is an example of a 10 bit mixing DAC that uses a 5 bit thermometer section 201 together with a 5 bit binary section 202, resulting in 36 mixer units in total. The mixers 2031-31 making up the thermometer unit 201 have an equal weighting, whereas the mixer units 20332-36 making up the binary section 202 have weights of ½, ¼, ⅛, 1/16 and 1/32 of the input RF signal respectively. Output signals from each of the mixer units 2031-36 are connected to a summation amplifier 204, and an output signal IFout is provided by the amplifier 204. An exemplary effective mixing waveform 301 resulting from such a mixer is illustrated in FIG. 3, the waveform being a quantized close approximation to a sinusoidal signal.
Compared to passive switching mixers, active Gilbert mixers tend to have somewhat inferior performance with regard to (1/f) noise and intermodulation.
An active mixer stage typically comprises a voltage to current converter 101 (also known as a transconductance amplifier) and a Gilbert mixer 102, as shown in FIG. 1. As the voltage to current converter 101 and mixer 102 are stacked, these blocks can share the same DC bias of 2×Is, which is determined by the two current sources 104a, 104b. The mixer IF output current can be converted into an output voltage using resistors or, more preferably, using a trans-impedance amplifier 401, an example of which is illustrated in the mixer 400 illustrated in FIG. 4, which incorporates the voltage to current converter 101 and Gilbert mixer 102 of FIG. 1. The voltage to current converter 101 may need to be operated at a higher DC bias current than the Gilbert mixer 102. This can be achieved by providing additional DC bias current by means of two optional additional current sources 403a, 403b connected between supply VDD and the drains of the voltage to current converter transistors. The trans-impedance amplifier 401 comprises a differential operational amplifier 403 with feedback resistors Rfb and capacitors Cfb connected across each input and output line.
A trans-impedance amplifier offers a low input impedance, thereby reducing voltage swings at the mixer output and contributing to improved signal handling. The use of a parallel resonance LC tank tuned to the IF frequency in order to convert the mixer output current into an output voltage is less attractive due to the size and associated cost of providing integrated inductors. To prevent suppression of the IF signal, and to prevent noise boosting of the trans-impedance amplifier 401, the DC bias provided by current sources 402a, 402b at the IF output terminals IF+ and IF− should have a high impedance at IF frequencies.
The pair of current sources 402a, 402b, providing currents ID, preferably operate in conjunction with a common mode control loop, as illustrated in the circuit 500 shown in FIG. 5. Different ways may be considered for implementing the DC bias currents, one of which is by means of a pair of degenerated PMOS transistors 601a, 601b, as illustrated in the circuit 600 shown in FIG. 6.
Noise generated by the DC bias current ID at the IF frequency will directly enter the IF amplifier 401 and thus affect the signal to noise ratio of the down-converted RF signal, IFout. For the case of a 10 bit mixing DAC the noise degradation by current sources ID was found to be unacceptably high. With a Mixer-IF amplifier overall voltage conversion gain of 17 dB and a single sideband Noise Figure of 21.4 dB (50Ω source) a (differential) DC current source noise density of 4.2 pA/√ Hz or better is required for a maximum of 0.4 dB noise contribution.
The above noise ceiling applies to the DC current source while it delivers 4 mA of DC current, as required by the 36 mixer cells that constitute the 10 bit mixing DAC shown in FIG. 2. The noise density of 4.2 pA/√ Hz equals the noise current produced in a 2×470Ω resistor. Lower noise requires a larger resistor, but a resistor of 470Ω carrying 4 mA already creates a DC voltage drop of 1.88V. The presence of a PMOS transistor 601a, 601b as shown in FIG. 6 will tend to add additional thermal and 1/f noise while further reducing the DC headroom available for the degeneration resistor 602a, 602b. Despite large area transistors and resistors, the simulated noise of such an implementation, as illustrated by the trace 701 in FIG. 7, is far beyond the desired noise requirement at 1 MHz, indicated by line 702. Occupying a reasonable 0.7 V DC headroom, the noise amounts to 16.2 pA/√ Hz in the flat part of the trace 701 (above around 10 MHz) and is 21.2 pA/· Hz at f=1 MHz.
What is needed therefore is a Gilbert-cell mixer having a DC load section that does not occupy considerable voltage headroom and yet does not add a significant amount of noise to the output.
The noise of the IF bias circuit 600 (FIG. 6) flowing directly into the IF amplifier 401 (FIG. 5) is distinct from the noise at IF frequencies generated by the bias current sources 104a, 104b (FIG. 1) in RF section 101 of the mixer 100. IF noise generated in the current Is is attenuated by commutation of the Gilbert mixer 102 inherent to the mixing process. Assuming a low frequency waveform with an accurate 50% duty cycle and a switching frequency fLO much higher than the IF frequency, this attenuation would be infinite if not for some circuit mismatch. In practice however, the noise contributed by Is can be neglected compared to the noise generated by the bias current source ID. The noise requirements of bias current source ID can be relaxed by applying commutation. Commutation is effective in attenuating the low frequency noise, but has the disadvantage of potentially introducing an unwanted spurious signal if applied with a frequency different from the mixing frequency.
It is an object of the invention to address one or more of the above mentioned problems.